The present invention relates to a switching power supply circuit including a voltage resonant converter.
As types of a so-called soft switching power supply that employs a resonant converter, a current resonant type and a voltage resonant type have been widely known. Currently, half-bridge connected current resonant converters formed of a two-transistor switching element have been widely employed since they can easily be put into practical use.
However, since characteristics of high-breakdown-voltage switching elements are currently being improved for example, problems about breakdown voltage associated with putting voltage resonant converters into practical use are being cleared up. Furthermore, it is known that a single-ended voltage resonant converter formed of one-transistor switching element is advantageous over a one-transistor current resonant forward converter with regard to input feedback noises and noise components of a DC output voltage line.
FIG. 9 illustrates one configuration example of a switching power supply circuit including a single-ended voltage resonant converter.
In the switching power supply circuit of FIG. 9, a voltage from a commercial alternating-current power supply AC is rectified and smoothed by a rectifying and smoothing circuit formed of a bridge rectifier circuit Di and a smoothing capacitor Ci, to thereby produce a rectified and smoothed voltage Ei as the voltage across the smoothing capacitor Ci.
The lines from the commercial power supply AC are provided with a noise filter that includes a pair of common mode choke coils CMC and two across-line capacitors CL, and removes common mode noises.
The rectified and smoothed voltage Ei is input to the voltage resonant converter as a DC input voltage. The voltage resonant converter has a single-ended configuration including one-transistor switching element Q1 as described above. The voltage resonant converter in this circuit is separately excited. Specifically, the switching element Q1 formed of a MOS-FET is switch-driven by an oscillation and drive circuit 2.
A body diode DD of the MOS-FET is connected in parallel to the switching element Q1. In addition, a primary-side parallel resonant capacitor Cr is connected in parallel to the channel between the source and drain of the switching element Q1.
The primary-side parallel resonant capacitor Cr and the leakage inductance L1 of a primary winding N1 in an isolation converter transformer PIT form a primary-side parallel resonant circuit (voltage resonant circuit). This primary-side parallel resonant circuit offers voltage resonant operation as the switching operation of the switching element Q1.
In order to switch-drive the switching element Q1, the oscillation and drive circuit 2 applies a gate voltage as a drive signal to the gate of the switching element Q1. Thus, the switching element Q1 implements switching operation with the switching frequency dependent upon the cycle of the drive signal.
The isolation converter transformer PIT transmits switching outputs from the switching element Q1 to the secondary side.
The isolation converter transformer PIT is constructed of an EE-core that is formed by combining E-cores composed of a ferrite material for example. Furthermore, the primary winding N1 and a secondary winding N2 are wound around the center magnetic leg of the EE-core, with the winding part being divided into the primary side and secondary side.
In addition, a gap with a length of about 1.0 mm is provided in the center magnetic leg of the EE-core in the isolation converter transformer PIT, so that a coupling coefficient k of about 0.80 to 0.85 is obtained between the primary side and the secondary side. When the coupling coefficient k has such a value, the coupling degree between the primary and secondary sides may be regarded as loose coupling, and thus it is difficult to obtain the saturation state. The value of the coupling coefficient k is a factor in setting the leakage inductance (L1).
One end of the primary winding N1 in the isolation converter transformer PIT is interposed between the switching element Q1 and the positive electrode of the smoothing capacitor Ci. Thus, the transmission of switching outputs from the switching element Q1 is allowed. In the secondary winding N2 of the isolation converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated.
In this circuit, a secondary-side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2. Thus, the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary-side parallel resonant capacitor C2 form a secondary-side parallel resonant circuit (voltage resonant circuit).
Furthermore, as shown in FIG. 9, connected to this secondary-side parallel resonant circuit are a rectifier diode Do1 and a smoothing capacitor Co, to thereby form a half-wave rectifier circuit. This half-wave rectifier circuit produces, as the voltage across the smoothing capacitor Co, a secondary-side DC output voltage Eo with the same level as that of an alternating voltage V2 induced in the secondary winding N2 (secondary-side parallel resonant circuit). The secondary-side DC output voltage Eo is supplied to a load, and is input to a control circuit 1 as a detected voltage for constant-voltage control.
The control circuit 1 detects the level of the secondary-side DC output voltage Eo input as a detected voltage, and then inputs the obtained detection output to the oscillation and drive circuit 2.
According to the level of the secondary-side DC output voltage Eo indicated by the input detection output, the oscillation and drive circuit 2 controls the switching operation of the switching element Q1 so that the secondary-side DC output voltage Eo is kept constant at a certain level. That is, the oscillation and drive circuit 2 produces and outputs a drive signal for achieving intended switching operation. Thus, stabilization control of the secondary-side DC output voltage Eo is achieved.
FIGS. 10A, 10B and 11 show results of experiments on the power supply circuit in FIG. 9. In the experiments, major parts of the power supply circuit of FIG. 9 were designed to have the following characteristics, as conditions for an AC input voltage VAC of 100 V, which corresponds to an AC 100 V-system input.
As the switching element Q1, a product of which breakdown voltage was 900 V was selected. As the secondary-side rectifier diode Do1, a product of which breakdown voltage was 600 V was selected.
The core of the isolation converter transformer PIT employed an EER-35 core, and a gap in the center magnetic leg thereof was designed to have a gap length of 1 mm. The numbers of turns T of the primary winding N1 and the secondary winding N2 were both set to 43 T. The coupling coefficient k of the isolation converter transformer PIT was set to 0.81.
The capacitances of the primary-side parallel resonant capacitor Cr and the secondary-side parallel resonant capacitor C2 were set to 6800 pF and 0.01 μF, respectively. Accordingly, the resonant frequency fo1 of the primary-side parallel resonant circuit was set to 175 kHz, and the resonant frequency fo2 of the secondary-side parallel resonant circuit was set to 164 kHz.
The rated level of the secondary-side DC output voltage Eo was 135 V. The allowable load power range was from the maximum load power Pomax of 200 W to the minimum load power Pomin of 0 W.
FIGS. 10A and 10B are waveform diagrams showing the operation of major parts in the power supply circuit in FIG. 9, while reflecting the switching cycle of the switching element Q1. FIG. 10A shows a switching voltage V1, a switching current IQ1, a primary winding current I1, a secondary winding voltage V2, a secondary winding current I2, and a secondary-side rectified current ID1, when the load power is the maximum load power Pomax of 200 W. FIG. 10B shows the switching voltage V1, the switching current IQ1, the primary winding current I1, the secondary winding voltage V2, the secondary winding current I2, and the secondary-side rectified current ID1, when the load power is the minimum load power Pomin of 0 W.
The switching voltage V1 is the voltage obtained across the switching element Q1. The switching voltage V1 has a waveform like those in FIGS. 10A and 10B. Specifically, the voltage level is at 0 level during the period TON when the switching element Q1 is in the on-state, while a sinusoidal voltage resonant pulse is obtained during the period TOFF when it is in the off-state. This voltage resonant pulse waveform of the switching voltage V1 indicates that the operation of the primary-side switching converter is voltage resonant operation.
The peak level of the voltage resonant pulse of the switching voltage V1 was 550 Vp when the load power was the maximum load power Pomax of 200 W and the input voltage VAC was 100 V (AC 100 V-system), and was 800 Vp when the load power was the maximum load power Pomax of 200 W and the input voltage VAC was 264 V (AC 200 V-system). In order to respond to these peak levels of the voltage resonant pulse, a product of which breakdown voltage was 900 V was used as the switching element Q1 as described above.
The switching current IQ1 is the current flowing through the switching element Q1 (and the body diode DD). During the period TOFF, the switching current IQ1 is at 0 level. During the period TON, the switching current IQ1 with a certain waveform like illustrated one is obtained. Specifically, at the time of turn on of the switching element Q1, the switching current IQ1 flows through the body diode DD in the forward direction thereof, and thus the switching current IQ1 has the negative polarity. After the turn on, the polarity is inverted and the switching current IQ1 flows between the drain and source of the switching element Q1. The current value increases with time until turn off of the switching element Q1. Therefore, the peak level of the switching current IQ1 is obtained at the timing of the turn off.
The primary winding current I1 flowing through the primary winding N1 is the current resulting from the synthesis between the current flowing as the switching current IQ1 during the period TON and the current flowing to the primary-side parallel resonant capacitor Cr during the period TOFF. Thus, the primary winding current I1 has a waveform like those shown in FIGS. 10A and 10B.
As the operation of the secondary-side rectifier circuit, the rectified current ID1 flows through the rectifier diode Do1 with having a certain waveform like that shown in FIG. 10A when the load power is the maximum load power of 200 W. Specifically, the peak level of the rectified current ID1 is obtained at the time of turn on of the rectifier diode Do1, and then the level gradually decreases toward 0 as shown in the waveform of FIG. 10A. During the period when the rectifier diode Do1 is in the off-state, the level of the rectifier diode Do1 is at 0. In contrast, when the load power is the minimum load power Pomin of 0 W, the current level is invariably at 0 even during the ON period of the rectifier diode Do1.
The secondary winding voltage V2 is obtained in the parallel circuit of the secondary winding N2 and the secondary-side parallel resonant capacitor C2. During the period when the secondary-side rectifier diode Do1 conducts, the secondary winding voltage V2 is clamped at the level of the secondary-side DC output voltage Eo. During the OFF period of the secondary-side rectifier diode Do1, the secondary winding voltage V2 shows a sinusoidal waveform of the negative polarity. The secondary winding current I2 flowing through the secondary winding N2 is the current resulting from the synthesis between the rectified current ID1 and the current flowing through the secondary-side parallel resonant circuit (N2 (L2)//C2). The secondary winding current I2 has the illustrated waveform for example.
FIG. 11 shows, as a function of load, the switching frequency fs, the ON period TON, the OFF period TOFF, and the AC to DC power conversion efficiency (ηAC→DC) of the power supply circuit shown in FIG. 9.
As for the AC to DC power conversion efficiency (ηAC→DC), an efficiency of 90% or more is achieved when the load power Po is in the range of 100 W to 200 W. It is known that a particularly single-ended voltage resonant converter, of which switching element Q1 is formed of one transistor, offers favorable power conversion efficiencies.
In addition, the switching frequency fs, the ON period TON, and the OFF period TOFF in FIG. 11 indicate the switching operation of the power supply circuit of FIG. 9 as the characteristic of constant-voltage control against load variation. In the circuit, the switching frequency fs is controlled so that the switching frequency increases as the load becomes lighter. As for the ON and OFF periods TON and TOFF, the period TOFF is almost constant independently of load variation. In contrast, the period TON is shortened as the load becomes lighter. That is, the power supply circuit in FIG. 9 varies and controls the switching frequency so as to shorten the ON period TON as the load becomes lighter while keeping the OFF period TOFF constant.
This variation control of the switching frequency allows variation of the inductive impedance that is due to the existence of the primary-side parallel resonant circuit and secondary-side parallel resonant circuit. This inductive impedance variation leads to a change of the amount of transmitted power from the primary side to the secondary side, and a change of the amount of transmitted power from the secondary-side parallel resonant circuit to a load. As a result, the level of the secondary-side DC output voltage Eo is varied. Thus, the secondary-side DC output voltage Eo is stabilized.
FIG. 12 schematically shows the constant-voltage control characteristic of the power supply circuit shown in FIG. 9, based on the relationship between the switching frequency fs (kHz) and the secondary-side DC output voltage Eo.
When the resonant frequencies of the primary-side and secondary-side parallel resonant circuits are defined as fo1 and fo2, respectively, the secondary-side parallel resonant frequency fo2 is lower than the primary-side parallel resonant frequency fo1 in the circuit of FIG. 9 as described above.
The characteristic curves in FIG. 12 are based on these resonant frequencies, and based on an assumption of constant-voltage control characteristics relating to the switching frequency fs, obtained for a certain constant AC input voltage VAC. Specifically, Characteristic curves A and B indicate the constant-voltage control characteristics obtained when the load power is the maximum load power Pomax and the minimum load power Pomin, respectively, based on the resonant impedance corresponding to the resonant frequency fo1 of the primary-side parallel resonant circuit. Characteristic curves C and D indicate the constant-voltage control characteristics obtained when the load power is the maximum load power Pomax and the minimum load power Pomin, respectively, based on the resonant impedance corresponding to the resonant frequency fo2 of the secondary-side parallel resonant circuit.
When a circuit includes a primary-side parallel resonant circuit and a secondary-side parallel resonant circuit like the circuit in FIG. 9, the center resonant frequency fo exists between the resonant frequencies fo1 and fo2. Characteristic curves E and F indicate the resonant impedance characteristics, based on the relationship between the center resonant frequency fo and the switching frequency fs, obtained when the load power is the maximum load power Pomax and the minimum load power Pomin, respectively.
In a voltage resonant converter including a secondary-side parallel resonant circuit, the level of the secondary-side DC output voltage Eo is determined depending on the resonant impedance characteristic with respect to the center resonant frequency fo, as a function of the switching frequency fs. Furthermore, the voltage resonant converter in FIG. 9 employs a lower-side control method, in which the switching frequency fs is varied and controlled in a frequency range lower than the center resonant frequency fo.
When, under the characteristics corresponding to the center resonant frequency fo, indicated by Characteristic curves E and F in FIG. 12, constant-voltage control in which the target value of the output voltage is the rated level of the secondary-side DC output voltage Eo (135 V, in the circuit of FIG. 9) is intended with use of switching frequency control based on lower-side control, the variable range of the switching frequency fs required for the constant-voltage control (requisite control range) is the range indicated by Δfs. That is, in the frequency range indicated by Δfs, the switching frequency is varied to a requisite value according to load variation. Thus, the secondary-side DC output voltage Eo is controlled so that it is kept at a rated level tg.
An example of the conventional power supply circuits is disclosed in Japanese Patent Laid-open No. 2000-152617.
In step with diversification of various electronic apparatuses, demands have been increasing for so-called wide-range compatible power supply circuits that operate in response both to commercial AC voltage inputs of the AC 100-V system and AC 200-V system.
The power supply circuit in FIG. 9 operates so as to stabilize the secondary-side DC output voltage Eo by switching frequency control as described above. The variable range (requisite control range) of the switching frequency required for the voltage stabilization is indicated by Δfs described in FIG. 12.
The power supply circuit of FIG. 9 is designed to respond to load variation of a comparatively wide variation range from 200 W to 0 W. In the power supply circuit of FIG. 9, an actual requisite control range of the switching frequency fs for this load variation condition is from 117.6 kHz to 208.3 kHz. That is, the range Δfs is 96.7 kHz, and this range is comparatively wide.
It is obvious that when the level of the AC input voltage VAC is changed, the level of the secondary-side DC output voltage Eo also varies. That is, the level of the secondary-side DC output voltage Eo varies depending on the level of the AC input voltage VAC.
Therefore, the level variation of the secondary-side DC output voltage Eo is larger when the AC input voltage varies in a wide range including both the AC 100-V system and AC 200-V system, compared with the level variation when the AC input voltage varies only in a single range of the AC 100-V system or AC 200-V system for example. In order to ensure constant-voltage control operation compatibly with the wide level variation of the secondary-side DC output voltage Eo, the requisite control range of the switching frequency needs to be widened from the above-described range of 117.6 kHz to 208.3 kHz so that higher frequencies are also covered.
However, in a present IC (the oscillation and drive circuit 2) for driving switching elements, the upper limit of a possible drive frequency is about 200 kHz. Even if an IC is developed so that an IC can drive switching elements with such a high frequency, the driving of switching elements with a high frequency leads to a significant decrease of the power conversion efficiency. It therefore is substantially impossible to put this IC into practical use for a power supply circuit.
As described above, it is very difficult to achieve a wide-range compatible power supply circuit by use of the configuration shown in FIG. 9 for example.
The power supply circuit shown in FIG. 9 includes a single-ended voltage resonant converter on its primary side. The power supply circuit with such a configuration tends to offer advantages for achieving a high power conversion efficiency as described above. However, in consideration of recent energy circumstances and environmental circumstances for example, electronic apparatuses have been required to have a further higher power conversion efficiency characteristic. Accordingly, a power supply circuit itself incorporated in an electronic apparatus has been required to have a further improved power conversion efficiency.